I don't have much knowledge about microcontrollers. In my project, I need to shift the sine wave. Here, I want to know, if I feed pure sine at port A pin 2. Then, will i get the shifted version of pure sine wave at port B pin 2 . will the following instruction work?
Inialise port A as input and port B as output
call delay
portb=porta
we can generate sine wave using DAC in microcontroller. But, as it is not perfect, it wont meet required conditions.
First of all the input needs to be to an ADC, and the output needs to be from a DAC (or a PWM with appropriate output filtering). It is not clear from your question that the pins you have chosen are appropriate for that.
If you are generating the sine from the DAC, why would you apply it to an input only to output it again? If you need two sine waves shifted in phase, why not simply generate calculated outputs from two DAC or PWM? Either way you need two analogue outputs, but that way you do not need any input. A PWM will need greater analogue filtering than a DAC and is likely to support lower bandwidth, but most microcontroller have more PWMs than DACs.
You cannot simply call a delay than copy port a to port b, that would be simply be a copy of a to b after a delay. You need to take samples from A and place then in a FIFO buffer, then apply the output of the FIFO to B. The length of the FIFO determines the delay.
A microcontroller is not an analogue device, you cannot put in an analogue signal on just any old pin and and transfer that signal to another pin. Most pins are digital GPIO, they except just two states representing 0 or 1. No matter what voltage you apply, it will be interpreted as either high or low.
Rather you will have to use an ADC input, sample at sufficiently high frequency, delay the samples through a FIFO, then apply the delayed samples to a DAC. Reconstruction of a "pure" sine wave from the quantized DAC output requires analogue filtering circuitry. With a filter cut-off lower than half the sampling rate you will recover a reasonably good representation of the original signal (which can be any signal with components below half the sampling frequency - it need not be a sine wave). If you do use a more complex signal, you will need to analogue filter the input to remove components above half the sampling rate to avoid aliasing.
It might be possible to do all that on one chip using a Cypress PSoC, since these are hybrid chips with reconfigurable analogue elements as well as a microcontroller.
Related
I have two USRP N210s connected together as a receiver and transmitter and I'm trying to send multi tone signals across the channel. However, I am finding extra frequency spikes mirrored about the centre frequency for when I send a signal with more than 2 tones.
I am using a low pass filter at the output with a cutoff frequency of 200kHz and the signal I am sending is constrained to 0-200kHz. I have an out of tree module that creates the multi tone signals that are evenly distributed within this bandwidth.
As I increase the number of tones the reflected frequency components become more and more prominent to the point at which I can barely correlate the input and output signals.
This is what the transmitted multi tone signals looks like
And here is the flowchart at the receiver end
The center frequency of the USRP source (receiver) is given by
uhd.tune_request(center_freq, rf_freq=(center_freq + lo_offset),rf_freq_policy=uhd.tune_request.POLICY_MANUAL)
which evaluates to 2.48GHz for which is the baseband frequency for the transmitting USRP
It may be something to do with the down conversion in the USRP or when GNURadio is actually sampling from the receiver with respect to this process.
Removing the LPF and connecting the FFT sink to the USRP source doesn't fix anything. The extra frequency spikes are still there (assuming the number of tones > 2)
You're not specifying how you transmit your tones into the RX port of your USRP, but the most likely thing to have happened here is that you didn't use sufficient attenuation between TX and RX – and you're saturating the RX amplifier chain (hopefully not damaging it).
By saturating, you drive them into nonlinear operation, which would lead to a broadband intermodulation.
Im currently creating a satellite ground station which will be used to control our cubesat in coming months. The modulation scheme used is GFSK and the baud rate is 9600. I have tried to run some tests by using a USRP board before I could try to communicate with the satellite by directly connecting the tx and rx blocks in the flowgraph.I was able to send and receive a png file using this flowgraph.
However, when I connect the tx and rx output to my USRP B210 TX/RX(transmission sink) and RX2(reception source) as shown below, I receive no data even though the two the source and sinks have been connected to each other carefully by RF cables with attenuators.
Below are the assumptions I took into account when I was making the second flowgraph. Please tell me if im on the right path.
Transmitter side : The packet decoder and GFSK mod blocks use 20 samples per symbol. Baud rate is 9600 and sample rate is 20*baud rate = 192K. Since the expected symbol rate by the satellite is baud_rate = 9600, I included a rational resampler and set UHD symbol rate to baud_rate. Is my logic correct?
GFSK mod and demod : For both of these blocks, I calculated sensitivity as S = Pi * Modulation_index/Samples_Per_Symbol. The default BT value of 0.5 is used. Are my calculations sound? Is there a link for to find documentation for GFSK blocks? My derivations are based on the GFSK python source code which is a poor substitute for documentation.
Packet Encoder/Decoder : The output of packet decoder is null even though the GFSK demod block give some kind of output which is rather meaningless. Is this normal? What is the meaning of the threshold variable and why its value is -1?
I'm a newbie in GNU Radio as well as GFSK in general. So please drop me any further references.
Thanks in advance.
Moses.
I was finally able to solve the problem. All I did was re-implementing the GFSK demod in GRC. If you go into source of gfsk.py, you will find out that the blocks used are Quadrature Demod --> M&M Clock recovery --> Binary slicer which can easily be connected in GRC directly. As Marcus suggested in my other thread, GFSK demodulation with Xlating filter in GNU Radio , I replaced the M&M Clock recovery block with PFB block. My flowgraph is shown below.
Even if I can not answer all of your questions, I provide below some thoughts:
When using hardware devices the Throttle MUST be removed from the flow-graph. The hardware device is now responsible for the rate limiting. Mixing hardware device and the Throttle block may break the real-time boundary of your flow-graph required by the device. Underflows or Overflows messages should be produced by the UHD driver in such a case.
Are you sure that the USRP can support the requested sampling rate? You may need also to change the master_clock_rate of the device, if the requested sampling rate is not an integer decimation of the clock. If this is not possible consider some kind of re-sampling.
EDIT: The B200 can not provide 192e3 sampling rate with the default clock. You can set the master_clock_rate at 19.2e6. The hardware will apply then the proper decimation. The master_clock_rate can be changed either by the device specific arguments or the Clock Rate field of the UHD Sink/Source blocks that presents in the latests GNU Radio versions.
I have a SCB 68A connector from National Instruments and I want to read out the open voltage from it. So I used the example code provided by National Instruments (https://decibel.ni.com/content/docs/DOC-28502):
I got 5 mV which is a reasonable value (I measured the noise signal with an oscilloscope). Now I want to read out the noise signal from few channels. So I sightly changed the VI (according to the documentation I need to create an array of channels and flatten them):
But now I read out approximately 200 mV on both channels (and one of them is the same as in the first VI). It doesn't make any sense.
What am I doing wrong?
I want the user to be able to choose the channels, so I can't just write "Dev1/ai0:4".
Edit: I'm using the DAQ 14.0.0.
Edit 2: 1) There is nothing connected to the deivce - I just want to read out the noise signal.
2) I'm using the connector in the MIO with the disabled temperature sensor mode (the default configuration).
You are observing charge injection from the DAQ device's multiplexer. Connect each aiN terminal to aignd and you will be able to measure the noise of the DAQ device.
Charge Injection
Most NI DAQ boards have a single analog to digital converter (ADC) and provide multiple input channels by using a multiplexer (MUX) to switch the input of the ADC to the different analog input terminals ai0, ai1, etc:
As NI explains, when the DAQ device's multiplexer moves from one channel to the next, it can introduce a small charge on each channel. Since the open channel does not have a path for this charge to dissipate, the voltage of the channel will increase. This can also cause the channel to rail, slowly floating up to the maximum input voltage (usually 10 V).
Characterizing Noise
You can determine the noise of each component in your system by:
Measuring the noise of the DAQ device
Measuring the noise of the DAQ device and terminal block
Subtracting the DAQ device noise (step 1) from the system noise (step 2)
When you're finished, the value from step 1 is the noise of the DAQ device, and the value from step 3 is the noise of the SCB-68.
To measure the noise of an electric path, there must be a complete circuit for the ADC to sample. For step 1, connect each aiN terminal to aignd and run your VI. For step 2, connect the terminal block to the DAQ device, disconnect the sensor, and connect the terminal block's channel terminals to its ground terminal and run your VI.
Minimizing Noise
In addition to charge injection, noise can be introduced to a DAQ system from several sources, including the environment. Open terminals act like small antennas and receive radiated energy from other electronics, lights, and the AC mains.
The link also outlines how to find and minimize noise, but the gist is:
Systematically identify the sources of the noise.
Remove sources of noise that aren't necessary for your measurements.
Depending on the nature and source of the remaining noise, use appropriate shielding, cabling, and terminal configuration.
Over-sample and average the signal.
Please have a look on the links below:
http://forums.ni.com/t5/Multifunction-DAQ/How-to-use-DAQmx-Read-to-measure-multiple-analog-channels/td-p/2620949
http://digital.ni.com/public.nsf/allkb/A3A05920BF915F1486256D210069BE49
There is the complete solution to your question.
Iam doing a project to recognize gestures by reading adc values in pic 16f73 using embedded c. Everything works fine while using single adc channel. When i use multiple channels, values are affected each other. is this a hardware error or software problem?
Probably. It's very likely to be one, or the other, or both. Split problem in half.
Eliminate one at a time. Scope/meter on both analog inputs. Change one input - does the other change too? If it does, there is a hardware issue at least. If not, it's software.
This is debugging 101.
It's a hardware effect, but not an error.
From the datasheet:
11.1 A/D Acquisition Requirements
For the A/D converter to meet its specified accuracy,
the charge holding capacitor (CHOLD) must be allowed
to fully charge to the input channel voltage level. The
analog input model is shown in Figure 11-2. The source
impedance (RS) and the internal sampling switch (RSS)
impedance directly affect the time required to charge
the capacitor CHOLD. The sampling switch (RSS)
impedance varies over the device voltage (VDD), see
Figure 11-2. The source impedance affects the offset
voltage at the analog input (due to pin leakage current).
The maximum recommended impedance for analog sources is 10 kΩ. After the analog input channel is
selected (changed), the acquisition period must pass
before the conversion can be started.
To calculate the minimum acquisition time, TACQ, see
the PICmicro™ Mid-Range MCU Family Reference
Manual (DS33023). In general, however, given a maximum source impedance of 10 kΩ and at a temperature
of 100°C, TACQ will be no more than 16 µsec.
It will likely be because you have high impedance sources driving all the ADC pins. When the multiplexer switches from one input to the next, any charge that is stored on the sampling capacitor of the ADC from the previous input will still be there.
If you drive each input with the output of a suitable op amp, when the ADC's multiplexer switches, the op amp is able to drive charge in or suck charge out of the sampling capacitor and the time needed for the new input you are reading can be significantly reduced. Plus, with this method you are not loading the voltage you are wanting to read.
If you cannot drive with a low impedance source, then ensure you have plenty of time for the new input's value to settle.
Hi Im using the following RF module
http://www.apogeekits.com/rf_receiver_module_rx433.htm
on an embedded board with the PIC16F628A. Sadly, I realized that the signal strength was in analog form and couldn't get any ideas to get the RSSI reading off the pin because well my PIC is digital DUH!.
My basic idea was
To get the RSSI value from my Receiver
Send it to the PIC
Link the PIC to a PC via RS232
Plot a graph of time vs RSSI of the receiver (so I can make out how close my TX is to my RX)
I thought it was bloody brilliant at first but ive hit a dead end here. Any ideas on getting the RSSI data to my PC from this receiver would be nice.
Thanks in Advance
You can get a PIC that has an integrated ADC for sampling the analog signal. Or, you can use an external ADC chip to do the conversion. You would connect that to your PIC using SPI or I2C.
The simplest thing to do is obviously to use a more appropriate microcontroller - one with an ADC! There are many (most), including PICs (though that wouldn't be my first choice).
Attaching an external SPI or I2C ADC might be a bit tedious since having no SPI or I2C on your part, you'd have to bit-bash it. If you do that, use an SPI part - its simpler. Your sample rate will suffer and may end-up being a bit jittery if you are not careful.
Another solution is to use a voltage controlled PWM, then use the timer input capture to time the pulse width. That will give you good regularity and potentially good resolution. You can get a chip (example) to do that, or grow your own. That last option requires a triangle wave input as well as the measured (control) voltage, but on the same site...
In a similar vein, you could use a low frequency VCO (example) and use the output to clock one of the timers, then using a second timer periodically sampling the first and reset it. The count will relate to the voltage, though not necessarily a linear relationship, linearisation could be none on the PIC or at the receiving PC - I'd go for the latter - your micro will suck at arithmetic (performance wise) - even integer arithmetic, especially if it involves division.